Power Electronics MCQ – 40 previous year questions with solutions

For a controlled half-wave battery charger, current can flow only after the source instantaneous voltage exceeds the battery emf and the SCR is fired. The conduction interval is bounded by the firing instant and the point where source voltage again falls to the battery voltage; evaluating the average current with the given source and load data gives the value in option A.

Source inductance prevents current from transferring instantaneously between rectifier devices. During the overlap angle μ, outgoing and incoming devices conduct simultaneously, reducing average DC output voltage. Key relation: Overlap introduces a commutation reactance drop; Vdc_actual < ideal Vdc.

Exam focus: More source inductance/current → more overlap. Common trap: Ignoring source inductance overestimates rectifier DC output.

A non-circulating-current dual converter prevents the two converter bridges from conducting simultaneously. Only one converter is enabled at a time; an interlocking/dead-time scheme blocks the other, so there is no closed path for circulating current.

A freewheeling diode gives inductive current a path when the main source/switch is removed. It clamps otherwise large inductive overvoltage, keeps current continuous and can improve efficiency and waveform quality in DC converter/rectifier circuits. Key relation: Inductor relation v=L di/dt explains the voltage spike if current path is interrupted.

Exam focus: Always provide a current path for stored inductive energy. Common trap: Freewheeling diodes are not limited to AC circuits.

Firing angle controls average DC voltage; the AC-side current is non-sinusoidal and its fundamental is displaced approximately by α. Vdc=(2√2×230/π)cos30°≈179.3 V. With R=10 Ω and highly inductive continuous current, Id≈17.93 A. Key relation: 1φ: Vdc=(2Vm/π)cosα≈0.9Vrms cosα; 3φ: Vdc≈1.35V_LL cosα.

Exam focus: Large load inductance → nearly constant DC current. Common trap: Distinguish displacement factor cosα from total power factor including distortion.

D — Remains unchanged. With continuous current each thyristor pair conducts for 180° until the next pair is fired; changing α only shifts the conduction interval, not its length.

An ideal half-wave rectifier passes one half-cycle of a sinusoidal input and blocks the other, creating pulsating DC. Vm=√2×100≈141.4 V and ideal half-wave average Vdc=Vm/π≈45.0 V. Key relation: V_m=√2 V_rms; V_DC=V_m/π; η_max≈40.6%.

Exam focus: Remember π in half-wave average; full-wave uses 2Vm/π. Common trap: Do not confuse diode ideality with 100% rectification efficiency.

D — Only during the positive half-cycle after triggering. The SCR is forward-biased only in the positive half-cycle and conducts from the firing angle α until the current falls to zero.

In three-phase bridges each thyristor conducts for 120 electrical degrees under continuous current. Vdc≈1.35×415×cos60°≈280.1 V. Displacement factor=cos60°=0.5. Key relation: 1φ: Vdc=(2Vm/π)cosα≈0.9Vrms cosα; 3φ: Vdc≈1.35V_LL cosα.

Exam focus: Large load inductance → nearly constant DC current. Common trap: Distinguish displacement factor cosα from total power factor including distortion.

B — AC to DC. A three-phase full-wave converter is a rectifier that converts three-phase AC input into DC output (six-pulse).

The primary is delta-connected, so its phase voltage is 1100 V. With a per-phase turns ratio of 5, the star-secondary phase voltage is 1100/5 = 220 V rms. The three-pulse rectifier's maximum load voltage is the phase-voltage peak, so I_max = √2×220/10 = √2×22 A.

For a single-pulse PWM waveform, the negative-half-cycle pulse is symmetrically located about 3π/2, just as the positive-half-cycle pulse is symmetric about π/2. This half-wave symmetry helps define the Fourier components.

For a single pulse of half-width d, the nth harmonic term contains sin(nd). To eliminate the nth harmonic, set sin(nd)=0; the non-trivial condition used in the options is nd = π.

In the linear region, modulation index controls the fundamental output amplitude. Raising switching frequency moves dominant harmonics upward but increases switching loss. Key relation: m_a=V_ref,peak/V_carrier,peak; linear SPWM typically m_a≤1.

Exam focus: m_a controls fundamental voltage in the linear region. Common trap: Over-modulation is not a clean linear extension of SPWM.

C — A triangular carrier wave. A sinusoidal reference is compared with a high-frequency triangular carrier; their crossings set the switching instants.

In 180° mode three devices conduct at a time; in 120° mode only two conduct and one phase is disconnected during each step. Waveform/fundamental comparison confirms 1. The 60° same-leg off interval of 120° mode contradicts 2, and switching-state neutral displacement contradicts 3. Key relation: Typical fundamental phase RMS: 180° mode ≈0.4502Vdc; 120° mode ≈0.3898Vdc. Thus 180° gives the higher fundamental component.

Exam focus: 120° mode: two switches ON, one phase floating. 180° mode: three switches ON. Common trap: Assuming a balanced load forces the floating neutral to stay at zero under a non-sinusoidal switching waveform.

Three-phase voltage-source inverters can be operated with 120° or 180° conduction sequences. In 120° conduction, each switch is on for 120° and only two devices conduct at a time, leaving one phase disconnected for intervals. Key relation: Conduction duration: 120°→two switches at a time; 180°→three switches at a time.

Exam focus: Track pole voltage and load-neutral voltage separately. Common trap: Balanced load does not force the floating neutral to the DC midpoint at every switching state.

In 180° conduction, three devices conduct at a time and the fundamental output is higher; the floating star neutral can shift relative to the DC midpoint. Full-bridge square-wave total RMS voltage is 48 V, so total load power=48²/10=230.4 W. Fundamental RMS=4Vdc/(π√2)≈43.2 V and fundamental power≈43.2²/10=186.6 W.

Exam focus: Track pole voltage and load-neutral voltage separately. Common trap: Balanced load does not force the floating neutral to the DC midpoint at every switching state.

D — Two. A half-bridge uses two controlled switches in series across a split DC supply; a full bridge needs four.

D — Pulse width. Single-pulse modulation uses one controlled pulse per half-cycle; varying its width varies the RMS output voltage.

B — Gate. A power MOSFET is a voltage-controlled device; the gate-source voltage switches it ON and OFF.

D — N-channel enhancement-mode MOSFET. Among the listed device types, it combines normally-off gate control with the low conduction resistance commonly sought in efficient power switching.

D — Three terminals, both carriers, current-controlled. A BJT has emitter, base and collector, conducts with both electrons and holes (bipolar) and is controlled by base current.

Power MOSFETs are majority-carrier, voltage-controlled devices with fast switching and a positive temperature coefficient of on-resistance that assists current sharing in parallel. IGBTs combine MOS-gate drive with bipolar conduction and exhibit tail current at turn-off, making them attractive at higher voltage/power but usually lower switching frequency than MOSFETs.

Exam focus: High-frequency low/medium voltage → MOSFET; higher voltage/power → often IGBT. Common trap: Both MOSFET and IGBT are not current-controlled devices.

A — It flows in reverse for a short time. Stored minority charge must be swept out, so the diode conducts a brief reverse current before it blocks (t_rr).

GaN wide-bandgap devices can switch at high frequency with low switching/conduction losses, enabling higher-efficiency and much smaller power converters than many conventional silicon implementations.

Commutation is the process of transferring current away from a conducting power semiconductor so that its current falls below the required holding value and the device turns off. In line-commutated circuits the AC source provides this naturally; in forced commutation an auxiliary network produces the required current/voltage condition.

D — Electrical isolation between control and power circuits. The opto-coupler passes the trigger signal as light, so the low-voltage control circuit is kept electrically separate from the high-voltage power circuit.

In the two-transistor equivalent of a GTO, anode shorts are introduced to alter regenerative action and improve turn-off behavior. The shorting path is represented between the emitter of T1 and the base region associated with T2, corresponding to the device's anode-short structure.

B — Direct light radiation. Light falling on the LASCR's gate-sensitive region generates carriers that turn it ON, giving optical isolation — useful in HVDC valves.

Latching current is the minimum anode current needed just after triggering so the device stays on when gate drive is removed. Holding current is the lower current threshold below which an already-conducting SCR turns off; latching current is normally higher, commonly a few times the holding current.

Exam focus: Gate pulse must persist until anode current exceeds latching current. Common trap: Do not substitute holding current for latching current during turn-on.

Thyristor protection separates dv/dt and di/dt problems. An RC snubber across the device slows the rise of off-state voltage and absorbs transient energy; a series inductance limits current rise during turn-on. Key relation: dv/dt protection → RC across switch; di/dt protection → series L.

Exam focus: Match the protection component to the stress being limited. Common trap: A capacitor alone can create severe discharge current without the snubber resistor.

Unlike an inverter, an AC voltage controller does not change AC to a new frequency; it controls the portion of each source cycle applied to the load. With phase control, delaying the firing angle reduces RMS output. Key relation: For a resistive single-phase phase-controlled load, Vrms depends on firing angle α; frequency at the load remains the source frequency.

Exam focus: AC controller → same frequency, variable RMS voltage. Common trap: Confusing an AC voltage controller with a frequency converter/inverter.

AC voltage controllers vary the rms AC voltage supplied to a load without changing the supply frequency. A classic application is resistive electric heating, where delivered power is smoothly controlled by phase-angle or integral-cycle control.

In an N-stage sequence controller, the supply voltage is divided across the staged secondary arrangement; each secondary is designed for a fraction of the total controlled voltage. The keyed relation is Vs/n per stage.

The load current is I = P/(V·pf) = 2000/(230×0.8) ≈ 10.87 A. With a safety factor of 2, the required SCR rms current rating is about 21.74 A.

An ideal boost converter in continuous conduction has an output voltage greater than or equal to input for 0≤D<1. The output capacitor supplies load current between energy-transfer intervals and reduces output ripple. Key relation: V_o=V_in/(1−D); ΔI_L≈V_in D/(Lf_s).

Exam focus: For an ideal boost, D=1−Vin/Vo. Common trap: A normal boost converter cannot produce Vo<Vin merely because D<0.5.

An ideal boost converter in continuous conduction has an output voltage greater than or equal to input for 0≤D<1. D=1−Vin/Vo=1−10/25=0.6. ΔIL=VinD/(Lf)=10×0.6/(100 μH×50 kHz)=1.2 A. Ideal average input/inductor current=P/Vin=25/10=2.5 A. Key relation: V_o=V_in/(1−D); ΔI_L≈V_in D/(Lf_s).

Exam focus: For an ideal boost, D=1−Vin/Vo. Common trap: A normal boost converter cannot produce Vo<Vin merely because D<0.5.

A single-phase midpoint cycloconverter uses two midpoint full-wave controlled converter groups—one positive and one negative. Each group requires two SCRs, giving a total of four SCRs.

SMPS units are compact and relatively tolerant of input-voltage variation, but their switching action can inject high-frequency ripple/noise and electromagnetic interference. Therefore 'SMPS has low input ripple' is the incorrect statement among the given choices.